Flyback eht and sawtooth current generator having a flyback period of at least sixth order

ABSTRACT

A flyback EHT and sawtooth current generator for television display apparatus including a transformer and switching means connected thereto which are conducting during a scan period and are non-conducting during a flyback period, and reactances whether parasitic or nuts connected to the transformer which together with the transformer constitute a network having a minimum of three prescribed resonant frequencies during the flyback so that a scan which is substantially without any oscillations and a flyback EHT pulse having a wide and flat peak is produced.

United States Patent [1 1 Pieters Oct. 30, 1973 [75] Inventor: RichardPieters, Emmasingel,

Eindhoven, Netherlands [73] Assignee: U.S. Philips Corporation, New

York, NY.

22 Filed: Apr. 18,1972

21 Appl. No.: 245,144

[30] Foreign Application Priority Data Moggre 315/27 R Boekhorst et315/27 GD Primary ExaminerBenjamin R. Padgett Assistant Examiner-J. M.Potenza AttarneyFrank R. Trifari [57] ABSTRACT A flyback EHT andsawtooth current generator for television display apparatus including atransformer and switching means connected thereto which are conductingduring a scan period and are nonconducting during a flyback period, andreactances whether parasitic or nuts connected to the transformer whichtogether with the transformer constitute a network having a minimum ofthree prescribed resonant frequencies during the flyback so that a scanwhich is substantially without any oscillations and a flyback EHT pulsehaving a wide and flat peak is produced.

6 Claims, 14 Drawing Figures Apr. 20, 1971 Netherlands 7105268 [52] US.Cl. 315/27 TD [51] Int. Cl. H01] 29/70 [58] Field of Search 315/27 R, 27TD, 315/276 D, 28, 29

[56] References Cited UNITED STATES PATENTS 3,500,116 3/1970 Rietveld315/27 TD I k l 18 SHEET 10F 9 PATENIEUncI 30 I915 PATENTEBHCI 30 msSHEET l 8F 9 PATENTEuummsn I Q 3,769,542

sum as; 9

Fig.7

FLYBACK EHT AND SAWTOOTH CURRENT GENERATOR HAVING A FLYBACK PERIOD OF ATLEAST SIXTH ORDER The invention relates to a flyback EHT and sawtoothcurrent generator particularly for television display apparatus,including switching means which are periodically non-conducting during aflyback period T and are conducting during a scan period T 'r and anetwork having input terminals connected to the switching means, saidnetwork comprising a transformer having at least one primary winding andpossibly one or more coils connected thereto, through which saidsawtooth current flows during the scan period, and a secondary windingto which a rectifier circuit is connected for generating said EHT fromthe voltage pulses occurring during the flyback period at the secondarywinding, said network having during the flyback period a first resonantfrequency f which is at least substantially equal to the expression inwhich K l and S is a correction factor which is equal to the relativereduction of the slope of the sawtooth current at the end of the scanperiod relative to this slope at the centre of the scan period, and asecond resonant frequency f; which is at least substantially equal tothe said expression for K 5.

[t is known from the Netherlands Patent Specification 88020 and from thebook Televisie by F.Kerkhof and W.Werner, third edition, chapter XIIl touse a transformer having one or more primary windings and a secondarywinding in flyback EHT and sawtooth current generators for televisiondisplay apparatus and to proportion the impedances of the circuitelements present such as the transformer windings,the leakage inductancebetween the primary windings and the secondary winding, the inductancesof the coils which are generally connected to a primary winding as wellas the parasitic and non-parasitic capacitances in such a manner thathardly any or no free oscillations occur in the secondary voltage duringthe scan period. Such free oscillations have the drawbacks that usefulenergy is lost, that this useless energy is mainly dissipated in thetransformer so that overheating of the transformer may occur and thatthe switching means conducting during the scan period may becomeprematurely nonconducting in case of large free oscillations.

The above-mentioned literature shows that said circuit constitutes a4th-order network during the flyback period with two resonantfrequencies and that the said free oscillations may be maintained low byproportioning the circuit elements in such a manner that these tworesonant frequencies f and f have very defined values which aredependent on the duration of the flyback period T and the duration ofthe scan period T r As is apparent from the expression given in thepreamblc, the optimum value Offer andf is also slightly dependent on theextent to which the slope of the sawtooth current flowing through thedeflection coils varies when the so-called S-correction of this currentis used.

In addition to the requirement that the scan period must be without freeoscillations a quite different requirement imposed for EHT and sawtoothcurrent generators is that the EHT generated varies as little aspossible when the EHT load varies. The generated EHT is generally usedin television display apparatus as an acceleration voltage for the beamcurrent in the display tube in which the average magnitude of the beamcurrent, which is dependent on the average luminance of the scenedisplayed, constitutes the EHT load. The extent to which the generatedEHT is dependent on its load is expressed by the so-called EHT R whichis the quotient between an EHT variation and the beam current variationcausing this EHT variation. As the EHT R, of a flyback EHT and sawtoothcurrent generator is lower, the EHT is less dependent on its load. Thisis of great importance because a high EHT R, (internal resistance)causes great power losses in the generator while a constant EHTfurthermore benefits the optimum adjustment of other magnitudesdependent on this EHT such as, for example, deflection and convergencein colour television display apparatus.

With the object of reducing the EHT R,- of a flyback EHT and a sawtoothcurrent generator while maintaining substantially no free oscillationsduring the scan period, the Applicant's Netherlands Patent application6714750 proposes to proportion the network in such a manner that thesecond frequency of the said TGSOLWTI frequencies is at leastsubstantially equal to the expression mentioned in the preamble for K 5(fifth harmonic tuning) while in addition the inductance (partly leakageinductance) and the capacitance (partly para sitic capacitance) presentbetween the primary windings and the secondary winding are chosen to besuch that the flyback pulse applied to the EHT rectifier has a peakwhich is as wide and as flat as possible. In fact,

a wide and flat peak of this pulse means that the EHT,

rectifier conducts during a large part of the flyback period and as aresult a reduction of the EHT R, of the circuit arrangement toapproximately 1 MD is obtained.

An object of the invention is to provide a flyback EHT and sawtoothcurrent generator in which a considerable further reduction of the EHTR, (to approximately 200 K0.) is possible while maintainingsubstantially no free oscillations and to this end the flyback EHT andsawtooth current generator according to the invention is characterizedin that said network includes further reactances increasing the order ofthe network during the flyback period to a minimum of 6 to such anextent that the network has a third resonant frequency f while is atleast substantially equal to the abovementioned expression for K 7.

It is essential for the invention than by a suitable addition ofreactances and/or the use of reactances already present a flyback periodnetwork of at least 6th order is obtained having at least three resonantfrequencies all of which substantially (i 10 percent) satisfy the givenexpression for K l, 5 and 7, respectively. Then substantial freedom ofoscillations during the scan period is obtained. In practice, someoscillations are found to occur predominantly as a result of theinevitable losses present in the network.

Both the flyback pulse occurring across the switching means and theflyback pulse applied to the rectifier now include three components, onehaving a frequency f one having a frequency f and one having a frequency f,,. The mutual amplitude ratios between these componentsdetermine the shape of the two pulses.

As is known a network having a plurality of resonant frequencies offixed values may be built up in many different manners; some examplesthereof will be given with reference to the Figures. It is found,however, that independent of the manner in which the network is built upthe mutual amplitude ratios of the three components of the flyback pulseoccurring across the switching means and hence the shape of this pulseis always dependent in the same manner on the three resonant frequenciesf ,f and f, as well as on the frequencies f and f (lying between fa .andfs and betweenf and f,,, respectively) at which the input im- Thecircuit arrangements according to the invention provide a possibility ofobtaining a secondary flyback pulse having a considerably wider andflatter peak than is the case in the above-mentioned known circuitarrangements. As already noted hereinbeforethe shape of the secondaryflyback pulse is determined by the mutual amplitude ratios between itscomponents. Furthermore it has been found that in whatever manner thenetwork is built up, these mutual amplitude ratios and hence the shapeof the secondary flyback pulse are always dependent in the same manneron the location of the possibly present zero transmission frequenciesrelative to the three resonant frequencies which are present and whichare determined by the expression given in the preamble. The zerotransmission frequencies of the network are those frequencies at whichthere is no transfer of energy from the input terminals to the rectifiercircuit in the network which is assumed to be free from losses. Unlikethe minimum impedance frequencies f, and f at the input terminals whichare always present and the first of which is always between f and f 6and the second is present betweenf and f,,, the number and the locationof the zero transmission frequencies is indefinite and is dependent onthe structure of the network.

When the network has only one zero transmission frequency f, or when asecond zero transmission frequency is so high that it does not havesubstantially any influence, a substantially flat and wide secondaryflyback pulse is obtained when this zero transmission frequency liesbetween f andf and this in such a manner that the ratio:

is between 9 percent and I7 percent.

It is to be noted that it is known to incorporate additional reactancesin a flyback EHT and sawtooth current generator. which reactancesincrease the order of the flyback network. However, these additionalreactances envisage objects which deviate entirely from the objects ofthe present invention. Therefore the proportioning of said extrareactances in the known generators is not such that these three resonantfrequencies are obtained.

The invention will be described in detail with reference to the figuresshown.

FIGS. 1, 4, 5 and 7 show different embodiments of a flyback EHT andsawtooth current generator according to the invention;

FIGS. 1a, lb, 10, 5a, 5b and 7a show different equivalent circuitdiagrams of the respective embodiments and FIGS. 2, 3, 6 and 8 showcharacteristic fields for further illustration of the proportioning ofthe different networks.

The embodiment of FIG. 1 shows a transformer having a primary winding 2,one or more auxiliary windings 3 rigidly coupled to the primary winding,a secondary winding 4 and a tertiary winding 5. A tap 6 on the primarywinding is connected to the positive terminal of a voltage supply source7 the negative terminal of which is connected to ground. The seriesarrangement of a plurality of deflection coils 9, a linearity correctorl0 and an S-correction capacitor 11 is arranged between a second tap 8and the lower end of the primary winding. The tap 8 and the lower end ofthe primary winding are located symmetrically relative to the tap 6 sothat this series arrangement is fed symmetrically relative to ground.

A transistor 12 operating as a switch is provided between the upper endof the primary winding and ground and a capacitor 13 is connected inparallel with this transistor. Said secondary winding 4 is connected toground at one end and at the other end to a rectifier circuit consistingof a rectifier 14 and a smoothing capacitor 15; the EHT generated by therectifier is applied to the acceleration anode of a television displaytube not further shown.

The lower end of the tertiary winding 5 is connected to the tap 6 of theprimary winding. A parallel LC- circuit which consists of an inductor l6and a capacitor 17 is arranged between the upper end of the primarywinding and the upper end of the tertiary winding.

Switching pulses which periodically cut off the transistor 12 at the endof each scan period are applied between the base and emitter oftransistor 12 through a separating transformer 18, a series inductance19 and a parallel diode 20. The transistor 12 is a so-called slowswitching transistor and the elements 19 and 20 are included so as toaccelerate the switching off of the transistor at the end of the scanperiod.

To explain the operation of the circuit arrangement described, FlG. lashows a first equivalent circuit diagram. In this Figure a switch SWdenotes the electronic switch constituted by transistor 12 and diode 20.The portion of the primary winding between the upper end of this windingand the tap 6 of FIG. 1 is denoted by 2'. The inductance of thedeflection coils 9 and the linearity corrector 10 is denoted by 21, buttransformed to said portion of the primary winding. 11 denotes the S-correction capacitor 11 likewise transformed to said portion. Theparasitic capacitance of the secondary winding as well as the inputcapacitance of the rectifier circuit is denoted by 22. The mutualmagnetic couplings between the primary, secondary and tertiarytransformer windings are denoted by M1, M2 and M3.

A further equivalent circuit diagram is shown in FIG. lb. in this Figurethe S-correction capacitor 11 is omitted because it has such a highvalue that it exerts substantially no influence during the flybackperiod. The capacitor 13 is directly provided across the inductor 21which is admissible because the impedance of the source 7 is very low.Furthermore the circuit diagram shows the magnetizing inductance 22 ofthe primary winding, the magnetizing inductance 23 of the tertiarywinding and the parasitic capacitance 24 of this winding, the inductor16 and the capacitor 17, the leakage inductance 25 between the primaryand the tertiary winding and in parallel therewith the parasiticcapacitance 26 between these windings, the leakage inductance 27 betweenthe tertiary and the secondary winding and in parallel therewith theparasitic capacitance 28 between these windings, the leakage inductance29 between the primary and the secondary winding and in paralleltherewith the parasitic capacitance 30 between these windings andfinally the capacitance 22 which represents the transform of thecapacitor 22 of FIG. la. [t is found that the capacitors 24, 28 and 30have such a low value in practice that they may be left out ofconsideration and that the inductor 23 has such a high value relative tothe inductor 16 that the inductor 23 may be displaced in parallel withthe inductor 22. The equivalent circuit diagram of FIG. 10 is obtainedwhen the inductors 21, 22 and 23 are combined to one inductor Ll, theinductors 25 and 116 to one inductor L3 and the capacitors l7 and 26 toone capacitor C3 and when furthermore the inductor 29 is denoted by L2,the inductor 27 by L4 and the capacitor 22 by C2 and when finally thecapacitor 13 is denoted by C1 in which also the parasitic capacitancesof the transistor 12, the primary winding, the deflection coils and thepossible auxiliary windings 3 are represented.

During the scan period switch SW is closed. The voltage E from voltagesupply source 7 is therefore present across capacitor C1 and also acrossinductor L1. As a result a (sawtooth) current linearly varying with timewill flow through the inductor Ll. When, as a result of a pulse appliedto the base electrode of transistor 12, switch SW is renderednon-conducting, free oscillations will occur in the network as a resultof the magnetic energy present in L1. These oscillations producepulsatory voltages V] and V2, the so-called flyback pulses acrosscapacitors C1 and C2, respectively. As soon as the flyback pulse acrossCl decreases to the value of the supply voltage E, that is to say, assoon the collector potential of transistor 12 becomes negative relativeto ground, the collector-base pn-junction of the transistor is in theforward direction and the next scan period commences. Switch SW in theequivalent circuit diagram of FIG. 10 therefore closes automatically assoon as the flyback voltage present across this switch becomes equal tozero.

It is to be noted that the sawtooth current in the circuit diagram ofFIG. 1 flows during the first part of the scan period through the diode20, the base-collector junction of the transistor and subsequentlythrough the transformer and the deflection coils to the voltage supplysource and thus feeds back energy to the voltage supply source. Sometime after the commencement of the scan period the base-emitterjunctionof the transistor is rendered conducting by means of the pulses appliedto the base electrode of the transistor so that during the second partof the flyback period the sawtooth current now reversed in polarity canflow from the voltage supply source through the transformer and thedeflection coils and subsequently through the collector electrode andthe emitter electrode of the transistor to ground, while the voltagesupply source supplies energy to the network.

It is to be ensured that during the scan period only a sawtooth currentflows through L1 and that free oscillations do not occur as a result ofelectrical or magnetical energy present in the inductors L2, L3 and L4and capacitors C2 and C3. Such a scan period without oscillations isobtained if it is ensured that the currents flowing through L2, L3 andL4 are all equal to zero during the entire scan period and thereforealso at the commencement and the end of the flyback period and that thevoltages across C3 and C2 are equal to zero and are equal to the batteryvoltage E, respectively.

To satisfy this condition the following two relations are to apply foreach resonant frequency a of the flyback period network, thus of thenetwork in case of an open switch SW:

a r K7T+ 211) 3 1 n liol In this relation 1' is the duration of theflyback period, K is each odd positive integer, i,, is the value of thesaw tooth current at the commencement of the flyback period, 1",, is thederivative with time of this current at the commencement of the flybackperiod and d) a is a phase angle. It is possible to eliminate qb fromthe two equations. Then a power series in 'r i'o/l}, is produced for ar. If this series is limited to the first two terms it is found that:

When a purely linear sawtooth current flows through L1 and hence throughthe deflection coils, there applies as an approximation: (ri,,/i,,) (2'r/T 'r) in which T- 'r is the duration of the scan period. When,however, the deflection current has a slightly S-shaped character as aresult of the S-corrcction capacitor of FIG. 1, which character isconventional in television display apparatus, there applies as anapproximation: (ri,,/i,,) (2 'r/TT) (1 A; S) in which S is the relativereduction of the slope of the deflection current at the end of the scanperiod relative to this slope in the centre of the scan period. Theabove-mentioned condition for arr then changes to:

ar= K 1r[l+(4/K 1r )(1'/T 'r) (l S)] withf a (oz/211') there followsthat f (K/2 T) [1 (l/K 17 ('r/Tr) (l S)] (III) As already notedhereinbefore the equations I and II and hence also equation III mustapply to each resonant frequency of the flyback network in order toobtain a scan without oscillations. In the circuit arrangementsaccording to the invention the flyback network is built up in such amanner that a network of the 6th order at a minimum is obtained havingat least three resonant frequenciesf ,f 6 and f and the porportioning ofthe network values is chosen in such a manner thatf a satisfies equationIII for K l,f for K 5 and f for K 7.

As is known (see for example the: book Theorie der Wisselstromen byProf.Ir.B.D.H.Tellegen, Vol. III particularly section 2, 351) the orderof a network is determined by the number of capacitor voltages which canbe given independently of each other and by the number of inductioncurrents which can be given independently of each other. In the networkof FIG. 1c all capacitor voltages and all induction currents can begiven independently of each other and the network therefore has theorder 7. However, there is one direct current solution namely in themesh formed by L2, L3 and L4. Since furthermore the network is supposedto be without resistors, the equation for the free oscillations of thenetwork is: p p +11 p 6 p y) O in which p is the differential operatortransformed in accordance with the Laplace arithmetic and a, e and yrepresent the three resonant frequencies of the flyback network,expressed in rad/sec.

A further consideration of the equivalent circuit diagram of FIG. lbshows that seven inductors are present therein whose currents can all begiven independently of each other; however, only three voltages of theseven capacitors present can be given independently of each other, forexample the voltages of the capacitors 13, 24 and 22, then the voltagesacross capacitors 17, 26, 28 and 30 are determined. The networktherefore has the order 10.. However, there are four direct currentmeshes present, namely a first mesh 21, 22, a second mesh 22, 16, 23, athird mesh 16, 25 and a fourth mesh 25, 27, 29. The equation for thefree oscillations therefore becomes:

p4 p2 2) p2 2) (p2 y2) The simplifications which are made at thechange-over from the circuit diagram of FIG. lb to that of FIG. 10 thushave not resulted in a reduction of the number of resonant frequencies.

The above given relations apply to a network in which energy is notdissipated. This is of cource not the case in practice because thenetwork elements are not completely without resistance and becauseenergy is drawn from the network through the rectifier circuit 14-15 andthrough the auxiliary windings 3. Therefore it will not be possible inpractice to realize a scan which is completely without any oscillation.However, since the electromagnetic energy circulating in the flybacknetwork is in practice considerably larger than the dissipated energy,the given relations remain approximately applicable and for a scan whichis sufficiently without oscillations the three resonant frequencies f fand f,, may therefore nOt deviate by more than 10% from the value foundin equation Ill.

When the input impedance of the network is determined as a function ofthe frequency, i.e. the impedance at the terminals to which theswitching means SW are connected, this input impedance will be at amaximum at the three resonant frequencies a, e and y. As is known afrequency is present between a and e and between 6 and y at which theinput impedance is at a minimum (in a network without resistors equal to0). These two zero impedance frequencies are hereinafter denoted by fand f and the associated angular frequencies are denoted by ,8 and ,8Thus there always applies that:

The flyback pulse which will be produced during the flyback period atthe input of the network has three sine functions, one for eachfrPquency, as a result of the three resonant frequencies of the network.When sin fl,

sin (yr-fl sin fl sin [6,,

+ A (IV) in which t represents the time during the flyback period hence0 s l s 1' and in which 4) a d) 6 and q), are the phase angles whichfollow from equation I which applies to each resonant frequency:

These values are thus found to be dependent only on the supply voltage Eand on the mutual location of the B, and B frequencies relative to a, eand y.

The shape of the flyback pulse is dependent on the mutual amplituderatios of the three respective sine oscillations. The amplitudes are,respectively, (A a/sin (b a), (A e/sin d) e) and (A /sin da Since inpractical cases d) a, d) e and 4),, are only small phase angles, sin(1)01, sin (1) e and sin 4),, may be replaced by lg d) a lg d) E and lg(1:, with which the amplitudes change to In view of the fact thatequation ll applies to each resonant frequency, a lg (b a e tgd) 6 ylg i/i For the ratio P between the amplitudes of the e and the aoscillations it is then found that P, A 5 ,A a and that for the ratio Pbetween the amplitudes of the Y and the a oscillations it is found thatP (yAy/ a A a When the above given expressions for A a A E and A aresubstituted therein, there follows:

P m wween e?" ml ausgwnw n (VII) P (B?y )(By FIG. 2 shows acharacteristics field in which horizontally B, varies from a to e andvertically B varies from 6 to y. The Figure shows lines of constant P,and constant P These lines clearly show that as the left top corner ofthe field is more and more approached the ratio P, increases; that is tosay, the amplitude of the 6 component relative to the amplitude of the acomponent increases. Likewise the amplitude of the y-component willincrease relative to that of the a-component as the left bottom cornerof the field is approached. The field furthermore shows a shaded area.In this area the amplitude of the e-component and/or the amplitude ofthe y-component is so large relative to the amplitude of the a-componentthat the flyback pulse (the voltage V,) occurring at cross capacitor C,becomes lower than the supply voltage -E already before the flybackperiod 1- given in equation I has elapsed. This results in thetransistor l2 conducting prematurely. The scan period then commenceswhile electrical and/or magnetical energy is still present in thenetwork elements which gives rise to free oscillation phenomena in thescan period. For a scan without oscillations it is therefore necessarythat:

l. the three resonant frequencies a, e and y of the network satisfyequation ill,

2. either switching means are used which remain nonconducting during theentire flyback period even when the voltage across the switching meansreverses its sign, or the B, and B frequencies are located in the areawhich is not shaped in FIG. 2. The latter condition may approximately beexpressed by the empirical relation:

The characteristics field and the shaded area of FIG. 2 are determinedby means ofa computer. In this case a, e and y are determined inaccordance with equation III with K l, 5 and 7, respectively, and at aflyback ratio 'r/T 0.18 and an S-correction factor S O. Modification ofthis flyback ratio and of the S-correction factor within limitsoccurring in practice results, however, hardly in a change of thepicture shown in FIG. 2.

It is furthermore to be noted that the above given equations andconditions are not limited to the network of FIG. 1c, but generallyapply to any network having three resonant frequencies, thusirrespective of the manner in which the network is actually composed ofthe different network elements. Of course the manner in which the a, e,y, ,8, and B frequencies are dependent on the separate network elementsis different for each network. lt is therefore hardly useful todetermine these dependences, also because the values of the differentnetwork elements are difficult to determine in practice due to theirfrequently entire or partly parasitic character. The five mentionedfrequencies a, e, y, ,8, and ,8, are, however, simply measurable towhich end the following method may be carried out. The arrangement inwhich the generator is included is switched off so that the voltagesupply source 7 does not provide voltage and the base connection oftransistor 112 is disconnected. A tone generator which covers therelevant frequency range is connected to the collector electrode of thetransistor through a sufficiently high impedance. Furthermore a voltagemeasuring instrument having a sufficiently high input impedance isconnected to the collector electrode, for example, an oscilloscope or avalve voltmeter. By varying the frequency of the tone generator thefrequencies are found at which the measured voltage is at a maximum,i.e. the resonant frequencies a, e and y, as well as the frequenciestherebetween at which the measured voltage is at a minimum, i.e. theminimum impedance frequencies Br and B2- As described in the preamble itis important for a low EHT R, that the flyback pulse V applied to therectifier circuit during the flyback period has a peak which is as wideand as flat as possible. Likewise as the flyback pulse V, the flybackpulse V consists of three components, one of the frequency a, one of thefrequency e and one of the frequency y. When a, e and y all satisfyequation III, we find for V sin (yr-(ii) sin l y (VIII) in which for B aB 6 and B,, there applies:

(l/ofi-l/afi) (l/ofi-ll/S'f) In these formulas E is a constant which isproportional to the supply voltage E and which is equal to E in thecircuit diagram of FIG. 10. 8 and 6, are the zero transmissionfrequencies of the network, i.e. those frequencies at which there is noenergy transmission in a network which is completely without resistorsbetween the input terminals of the network and the output terminals towhich the rectifier circuit is connected. Due to the inevitable lossessome energy transmission will take place at these frequencies inpractice.

The above given expressions for B B E and 8,,

generally apply to each network having three resonant frequencies whichhas a scan period without oscillations in the manner described. However,unlike the frequencies ,8, and B both of which are always present andthe lowest ([3,) of which is always between a and e and the highest([3,) is always between 6 and y both zero transmission frequencies maynot always be present. In many circuit arrangements according to theinvention only one zero transmission frequency (5,) will be present orthe second zero transmission frequency (8,) is so high that it may beleft out of consideration. Thus in the equivalent circuit diagram ofFIG. 10 only one zero transmission frequency is present namely thefrequency at which the two-pole formed by L2, L3, C3 and L4 is inparallel resonance, hence 8,, (L2 L3 L4)/C3L3 (L2 L4).

The equations for B a B i and B then change to:

When S, represents the ratio between the amplitude of the e-componentand the amplitude of the a-component and when 5 represents the ratiobetween the amplitude of the y-component and the amplitude of thea-component, there applies in a corresponding manner as for the ratios Pand P of the flyback pulse V, that:

S, B /aBa) and S B laBa) With the given expressions for Ba, B and 8,, itis found for the case where there is only one zero transmissionfrequency:

When the ratio S is determined in accordance with values of a, e and ylaid down in accordance with equation III and by a given choice of 8 theratio S is of course also determined. The relationship between S, and Sis as follows:

a+eS +yS =O XII It will be evident from the foregoing that when thenetwork has only one zero transmission frequency, the location of thiszero transmission frequency relative to a, e and y is decisive for theshape of the flyback pulse V It follows from a further consideration ofthe three components which jointly constitute the flyback pulse V thatwhen a, e and y satisfy equation III with K l, 5 and 7 respectively, itis necessary for a pulse having a wide and flat peak that both S and Sare negative. This is possible only when 8 lies between G and y while apossible second zero transmission frequency 8 must be higher than y.

For an acceptable shape of the flyback pulse the zero transmissionfrequency 8,, will lie between 6 and y in such a manner that S, isbetween 0.09 and 0.l7 while an optimum flyback pulse is obtained at an Sof approximately O. 14.

Unlike the frequencies ,8 and B which as stated above can be determinedin a simple manner in a practical generator this is considerably moredifficult for the zero transmission frequencies. This is mainly a resultfrom the fact that in practical circuit arrangements the impedance levelat the output terminals coupled to the rectifier is very high. Theparasitic capacitances of measuring equipment connected to the terminalsthen cause a considerable unwanted variation of the proportioning of thenetwork.

When, however, the structure of the network, in this case the structureof the equivalent circuit diagram is known, it is possible in many casesto determine the relationship between the zero transmission frequenciesand the zero impedance frequencies B, and B and thus to determine thelocation of the zero transmission frequencies through measurement of ,B,and [3 Thus for the equivalent circuit diagram of FIG. 10 there appliesthe following relation FIG. 3 shows with the aid of the above-mentionedequation the field of FIG. 2, but in this case with lines of constant 8Since the frequency 8,, also determines the ratios 8, and S and hencethe shape of the flyback pulse V these are also lines where the flybackpulse V has a constant shape. The reference 0 denotes the line at whichthe ratio S 0.l7 while S O. 14 is given on line b and S 0.09 on line 0.This field also shows the shaded area already mentioned with referenceto FIG. 2. In case of an EHT and sawtooth current generator for which atleast the equivalent circuit diagram of FIG. 10 substantially appliesthe frequencies B, and B will therefore be chosen between the lines aand c of FIG. 3 and preferably in the non-shaded area.

In the embodiment of FIG. 4 corresponding elements have the samereference numerals as those in FIG. 1. Unlike FIG. 1 the primary windingconsists of two identical halves 2a and 2b while the voltage supplysource 7 is arranged between the two halves. The series arrangement ofS-capacitor l1, linearity corrector 10 and deflection coils 9 issymmetrical relative to ground between the high end of the upper half 2aand the low end of the lower half 2b. The transistor 12 and thecapacitor 13 are included between taps 32 and 33 of the two primaryhalves which are arranged symmetrically relative to ground. The tertiarywinding 5 is also connected between these taps and this through theLC-circuit l6 and 17 and through a direct voltage isolation capacitor 31of large value. The advantage of the network of FIG. 4 relative to thatof FIG. 1 is that not only the series arrangement of the elements 9, 10and 11 but also the primary winding, the transistor 12, the capacitor 13and substantially the tertiary winding 5 are located symmetricallyrelative to ground; this yields a considerable reduction of theparasitic radiation of the generator. The equivalent circuit diagram ofthe flyback network of this embodiment is the same as shown in FIG. 10so that also the above described phenomena are the same.

It is to be noted that in the embodiments of FIG. 1 and FIG. 4 theleakage inductance between the primary and secondary windings do notplay a principal role. In fact, if in the equivalent diagram of FIG. 1cthe inductor L2 representing this leakage is omitted, a network stillremains having three resonant frequencies for which all considerationsdescribed above apply. The leakage between the secondary and thetertiary winding represented in FIG. 10 by the inductor L4 plays,however, a very essential role. In fact, when the coupling betweensecondary and tertiary winding is fight in such a manner that L4 in theequivalent circuit diagram can be replaced by a short circuit, thevoltages across capacitors C1, C2 and C3 can no longer be givenindependently of each other and therefore a 5th order network remainshaving one direct current solution (through L2 and L3). The network thenonly has two resonant frequencies.

The embodiment of FIG. 5 deviates from the embodiment of FIG. 1 in thatthe parallel arrangement of the inductor l6 and the capacitor 17 islocated between the high end of the primary winding 2 at one end and thecapacitor 13 and the collector of the transistor 12 at the other end. Animportant advantage of this circuit relative to the circuit arrangementof FIGS. 1 and 4 is that the tertiary transformer winding 5 can beomitted. For the arrangement of FIG. 5 a simplified equivalent circuitdiagram can be set up as is shown in FIG. 5a. In this FIG. Cl representsthe capacitor 13 and the output capacitance of the transistor, L3 and C3represent the inductor l6 and the capacitor 17, respectively, Llrepresents the transformed inductance of the deflection coils 9 and thelinearity corrector 10 as well as the magnetizing inductance of thetransformer, L2 represents the leakage inductance between the primarywinding and the secondary winding and C2 represents the parasiticcapacitance of the secondary winding and the input capacitance of therectifier circuit all transformed to the primary side. The network ofFIG. 5a is a 6th order network having three resonant frequencies a, eand y. For a scan period without oscillations there must not apply bothat the beginning and at the end of the flyback that the current throughL3 and the voltage across C3 are zero because the sawtooth currentflowing through Ll also flows through L3. The conditions for a scanperiod without oscillations therefore are that at the commencement andend of the flyback:

l. the current through L2 I 2. the voltage across L2 0, hence thevoltage across C2 is equal to the voltage across LI 3. the currentthrough L3 is equal to the current through Lll, hence the currentthrough C3 is equal to O 4. the ratio of the voltage across C3 and L3and the voltage across L1 is the same as the ratio between the inductorL3 and the inductor L1.

It is found that these four requirements are satisfied when the abovegiven equations l and II are satisfied, while i and 1" again representthe current through L1 and the derivative of this current, respectivelyat the end of the scan period. The equations I and II and hence theequation III derived therefrom are found to be generally applicable asconditions for a scan without oscillations. Also equations IV, VI andVII which relate to the shape of the flyback pulse occurring acrossswitch SW are found to be generally applicable for a flyback networkhaving three resonant frequencies. Finally also the equations VIII andIX for any such network are found to be applicable. Since the equivalentcircuit diagram of FIG. 5a has only one zero transmis sion frequency,namely the frequency at which L3 and C3 are in resonance, the equationsX, XI and XII also apply to this equivalent circuit diagram.

The relationship between the zero transmission frequencies and thefrequencies a, e,,, B, and 8 which was given for the circuit diagram ofFIG. 10 by equation XIII is, however, different. This relationship isobtained for the equivalent circuit diagram of FIG. 5a by choosing for 8the highest root from the following square equation in 8 When the linesof constant 8,, and hence the lines of constant S and S are determinedwith the aid of this equation in the ,B ,8 field, the field of FIG. 6 isfound. As can be seen the lines for S 0.09, O.l4 and 0.l7 vary in thenon-shaded area of this field in approximately the same manner as in thefield of FIG. 3. For an optimum secondary flyback pulse the B and Bfrequencies will therefore be located in an approximately correspondingmanner.

It is to be noted that since V represents the voltage across thesecondary winding 4, the equivalent circuit diagram of FIG. 50 would becorrect when the low end of the secondary winding were directlyconnected to ground. However, since this low end is connected to thehigh end of capacitor 13 the voltage V V NV is actually applied to therectifier in which N is the lid transformation ratio of the transformer.The equivalent circuit diagram of FIG. 5 then becomes actually thediagram shown in FIG. 5b in which T is an ideal transformer having atransformation factor N because the magnetizing inductance and theleakage inductance of the transformer are already accounted for in L1and L2. It will be evident that when N is large relative to l, theflyback pulse V substantially does not exert any influence on the outputpulse V .and the output pulse will therefore have substantially the sameshape as the pulse V When N is not large relative to I such as mayoccur, for example, in valve circuits the share of V in V will have tobe taken into account. It is found that even now the equations VIII andIX maintain their general applicability if in the left member of VIII Vinstead of V is read and if 8 and 8 are defined as those zerotransmission frequencies at which no energy transmission occurs to theactual output terminals, hence those frequencies at which NV V O. Thecircuit diagram of FIG. 511 has two such zero transmission frequencies.When N is large relative to l, the first zero transmission frequency (8is located very closely to the resonant frequency of L3 and. C3 whilethe second (8 is considerably much higher.

Corresponding considerations apply when, for example, in the embodimentsof FIG. 1 and FIG. 4 the low end of the secondary winding 4 is connectedto the high end of the tertiary winding 5 instead of to ground. The twowindings 4 and 5 then together constitute substan tially one windinghaving a tap leading to the LC-circuit 16-17.

A drawback of the embodiment of FIG. 5 is the following. As a result ofthe losses present in the flyback network and tolerance deviations ascan which is completely without any oscillation cannot be realized inpractice. In the embodiments of FIG. l and FIG. 4 the series arrangementof deflection coils, linearity corrector and S-capacitor is arrangedthrough the primary winding substantially directly across switchingtransistor 12. This is still more evident from the equivalent circuitdiagram of FIG. 1c where the inductor Ll is directly connected to switchSW. Scan oscillations which occur in the section L2, C2, L3, C3 and L4cannot reach the deflection coils (Ll) because Ll is short circuitedthrough the now conducting switch SW and through the voltage supplysource. In the embodiment of FIG. 5 this is, however, not the case; scanoscillations which occur across the resonant circuit L3, C3 are alsopresent across L1 and hence across the deflection coils. This causesunwanted modulation of the deflection in the display tube of thetelevision apparatus.

While obviating a tertiary transformer winding this drawback is notpresent in the embodiment of FIG. 7. This embodiment has the samesymmetrical structure as that of FIG. 4. The parallel arrangement ofinductor l6 and capacitor 17 is, however, arranged in series withcapacitor 13 while the tertiary winding 5 is omitted. FIG. 7a shows theequivalent circuit diagram having the most important network elements.As can be seen the flyback network is of the 6th order because sixinductors and capacitors are present and because all capacitor voltagesand induction currents can be given independently of each other. Inaddition, there are no direct current or direct voltage solutionspresent so that the 6th order network has again three resonantfrequencies a, e and y for which all given relations I to IX inclusiveapply. The network has one zero transmission frequency 8 namely thefrequency at which the twopole constituted by C1, C3 and L3 is in seriesresonance.

8 2 1/L3 (C1 C3),

so that also the equations X, XI and XII apply.

The relationship between 8 and the frequencies ,8, and ,8 is very simplein this case because the series resonant frequency of the dipole C1, C3,L3 is also one of the two zero impedance frequencies at the inputterminals, hence 8, B, or 6 [3 Since, as stated above, a flyback pulse Vhaving a wide and flat peak necessitates that 8,, lies between 6 and yand since furthermore B, is never between e and y and B is alwaysbetween a and y it follows that for 8 there applies that 8 [3 The linesof constant 8 and hence of constant S, and S in the B, B field aretherefore horizontal straight lines. In FIG. 8 this field is shown withthe position of the lines in which S 0.09, O.I4 and 0.17 respectively.

It is known from the network theory that the twopole constituted by C1,C3 and L3 may be replaced by an equivalent network comprising the seriesarrangement of L4 and Cl and the capacitor C3 in parallel with thisseries arrangement. The values of the components are then to be modifiedaccordingly.

Instead of being arranged in series with the capacitor 13, as shown inFIG. 7, the circuit 16-17 may be arranged, for example, in series withthe deflection coils in which the B ,6 field of FIG. 8 continues toapply. Such a circuit arrangement has, however, the drawback thatpossible remaining scan oscillations occur across the deflection coils.

As described hereinbefore it is impossible in practice to give completeand exact propertioning measures for the respective circuit elementswhich lead to the abovegiven frequency values. In practice the correctproportioning will be found by variation of the different circuitelements; namely by the choice of the number of turns of the transformerwindings; the manner of winding these windings so that the parasiticcapacitance as well as the mutual leakage thereof can be influencedwithin given limits, the choice of the various taps as well as theinductance and capacitances of the reactances connected to thetransformer. The following considerations may be useful for thispurpose.

I. If in all embodiments shown the parallel LC-circuit I6-I7 isshort-circuited, a network is always obtained having only two relevantresonant frequencies and of which the first is substantially equal to aand the second lies between 6 and y. The network having a shortcircuitedLC-circuit will thus firstly be proportioned in such a manner that thefirst of the two occurring resonant frequencies is approximately equalto a and the second lies between e and y; subsequently the LC- circuitis provided which is controlled in such a manner that a scan withoutoscillations as well as a flyback pulse V having a wide and flat peak isproduced.

2. The frequency a will generally be equal to l/ \l LI (C1 C2). Thisfrequency is thus determined in a first approximation by the inductanceof the deflection coils and the linearity corrector, by the location ofthe tap on the primary winding to which these elements are connected aswell as by the sum of the primary tuning capacitor 13 and the parasiticcapacitance of the secondary winding and of the rectifier circuit whichcapacitance is multiplied by the square value of the transformationratio of the transformer.

3. The second resonant frequency mentioned under item I and occurringwhen the LC-circuit is shortcircuited is determined to a considerableextent by the series arrangement ofCl and C2 and by the leakageinductance (L2 parallel to L4 in FIG. 1c and L2 in FIG. 5a 5b and 7a).Since this second resonant frequency must be between the ultimatelydesired values of e and y and since these are comparatively highfrequencies, this means that the leakage inductance between the primaryand secondary winding (in FIGS. 1 and 4 mainly the leakage inductancebetween tertiary and secondary windings) and the capacitance of thesecondary winding are to be comparatively low. In the embodiments ofFIGS. 1 and 4 the secondary and tertiary windings therefore bepreferably wound across each other on the transformer core while in theembodiments of FIGS. 5 and 7 this is the case with the primary and thesecondary windings. The parasitic capacitance of the secondary windingmay be maintained low by winding this winding in a narrow and highmanner; however, this measure increases the leakage inductance with theother windings so that a comprise is to be found. When a voltagemultiplying rectifier circuit is used, the num ber of turns of thesecondary winding and hence both the leakage inductance and theparasitic capacitance of this winding is lower. Proportioning thenbecomes considerably simpler. A drawback of such a voltage multiplieris, however, that the EHT R, is increased thereby.

Finally the following is to be noted. In the generators according to theinvention the three resonant frequencies a, e and y of the flybacknetwork at least substantially satisfy equation III for K being equal toI, 5 and 7, respectively. In practical cases the lowest resonantfrequency a will always satisfy equation III for K I. However, itremains possible for the three resonant frequencies to satisfy equationIII for K being equal to, for example, I, 3 and 5, respectively, or I, 3and 7, respectively. A scan without oscillations can then also berealized. It is, however, found that for a wide and flat secondaryflyback pulse in the cases I, 3, 5 and I, 3, 7 the lowest zerotransmission frequency must be considerably much lower than in the caseI, 5, 7, namely lower than the second resonant frequency. In practice,this leads to circuit arrangements which cannot be realized. A I, 3, 5generator is, for example, possible with a scan which is sufficientlywithout oscillations but in which the zero transmission frequency(frequencies) is(are) anything but optimum. The secondary flyback pulseconsequently has a much less wide peak as in the circuit arrangementsaccording to the invention. The EHT R, will hardly be lower in such acircuit arrangement than in the known circuit arrangements which haveonly two resonant frequencies.

Theoretically it is possible to make a network of at least 8th orderhaving four or more resonant frequencies with the aid of an additionalnumber of reactances which are either or not parasitic, while the fourthresonant frequency satisfies equation III for K being, for example,equal to 9 or I 1. It is found, however, that such a high resonantfrequency hardly exerts any influence on the shape of the flyback pulsesand in addition cannot be handled in practice because the inductancesand/or capacitances which lead to these extra resonant frequencies arevery low.

What is claimed is:

1. A flyback EHT and sawtooth current generator particularly fortelevision display apparatus including switching means which areperiodically nonconducting during a flyback period 1- and are conductingduring a scan period T 1- and a network having input terminals connectedto the switching means, the network compricing a transformer having atleast one primary winding and possibly one or more coils connectedthereto through which said sawtooth current flows during the scanperiod, and a secondary winding to which a rectifier circuit isconnected for generating said EHT from the voltage pulses occurringduring the flyback period at the secondary winding, said network, duringthe flyback period, having a first resonant frequencyf which is at leastsubstantially equal to the expression wherein K l and S is a correctionfactor which is equal to the relative reduction of the slope of thesawtooth current at the end of the scan period relative to this slope inthe middle of the scan period, and a second resonant frequency f G whichis at least substantially equal to the said expression for K 5,characterized in that in said network further reactances are presentwhich increase the order of the network during the flyback period to aminimum of 6 in such a manner that the network has a third resonantfrequency f,, which is at least substantially equal to theabove-mentioned expression for K 7.

2. A flyback EHT and sawtooth current generator as claimed in claim 1,characterized in that for the two frequencies f and f located between fandf and between f and f,,, respectively, at which the impedance at theinput terminals of the network is at a minimum there applies that 703-121 /f 'fa 1,46 (f2"fe fu fe 3. A flyback EHT and sawtooth currentgenerator as claimed in claim 1, characterized in that said network hasat least one zero transmission frequency in which the energy transferfrom the input terminals to the rec- 'tifier circuit is at leastsubstantially zero, said zero transmission frequency f being locatedbetween f and f in such a manner that the ratio:

is larger than 9 percent and is smaller than 17 percent.

4, A flyback EHT and sawtooth. current generator as claimed in claim l,in which the transformer has a tertiary winding and in which theswitching means and a tuning capacitor are connected to the primarywinding, characterized in that the tertiary winding is connected througha parallel LC-circuit to the primary winding and that the network. atleast due to the inductance of said coils, the capacitance of the tuningcapacitor, the parasitic capacitance of the secondary winding and therectifier circuit, the inductance and the capacitance of said parallelLC-circuit and the leakage inductance between the secondary and tertiarywinding constitutes a network of a minimum 6th order with the said threeresonant frequencies.

5. A flyback EHT and sawtooth current generator as claimed in claim 1 inwhich a tuning capacitor is connected through a connection to theprimary winding of the transformer, characterized in that a parallel LC-circuit is included in said connection and that the network, at leastdue to the inductance of said coils, the capacitance of the tuningcapacitor, the parasitic capacitance of the secondary winding and therectifier circuit, the inductance and the capacitance of said parallelLC-circuit and the leakage inductance between the primary and thesecondary winding constitutes a network ofa minimum of 6th order withthe said three resonant frequencies.

6. A flyback EHT and sawtooth current generator as claimed in claim 5,characterized in that the switching means are arranged across the seriesarrangement of the said tuning capacitor and the parallel LC-circuit.

Patent No. 3,769,542 Dated October 30, 1973 Inventor (s) RICHARD PIETERS It is certified that error appears in the above-identified patentand that said Letters Patent are hereby corrected as shown below:

- '1 col. 8, lines 16, 39, 40, 41 & 42, change "9!- to ,O

lines 12; 39, 4o, 41 a 42, change we to 1 lines 20, 23 a 39, change "Ad"to A lines 20, 27 & 39, change "A5" to A line 17, change "S W to 5 r 2 2line 18, change "12" to 7 col. 9, line 27, after "B insert a minus col.10, lines 10 & 11, change Kidto 31 change "91C to $5 (both occurrences)line 54, after "11/; 2" insert v a parenthesis col. 11, line 6, cancel"y y" and insert yBY IN THE CLAIMS col. 17, line S, cancel ".compricing"and insert comprising Signed and sealed this 17th day of September 1974.

(SEAL) Attest:

McCOY M. GIBSON JR. C. MARSHLL DANN Attesting Officer Commissioner ofPatents

1. A flyback EHT and sawtooth current generator particularly for television display apparatus including switching means which are periodically non-conducting during a flyback period Tau and are conducting during a scan period T - Tau and a network having input terminals connected to the switching means, the network compricing a transformer having at least one primary winding and possibly one or more coils connected thereto through which said sawtooth current flows during the scan period, and a secondary winding to which a rectifier circuit is connected for generating said EHT from the voltage pulses occurring during the flyback period at the secondary winding, said network, during the flyback period, having a first resonant frequency f which is at least substantially equal to the expression K/2 Tau (1 + (4/K2 pi 2) ( Tau /T - Tau ) (1 - 2/3 S)) wherein K 1 and S is a correction factor which is equal to the relative reduction of the slope of the sawtooth current at the end of the scan period relative to this slope in the middle of the scan period, and a second resonant frequency f which is at least substantially equal to the said expression for K 5, characterized in that in said network further reactances are present which increase the order of the network during the flyback period to a minimum of 6 in such a manner that the network has a third resonant frequency fy which is at least substantially equal to the above-mentioned expression for K
 7. 2. A flyback EHT and sawtooth current generator as claimed in claim 1, characterized in that for the two frequencies f1 and f2 located between f and f and between f and fy, respectively, at which the impedance at the input terminals of the network is at a minimum there applies that 4,61 (f1 - f /f - f ) - 1,46 > (1,36 (f2 - f /fy - f ) - 1)2.
 3. A flyback EHT and sawtooth current generator as claimed in claim 1, characterized in that said network has at least one zero transmission frequency in which the energy transfer from the input terminals to the rectifier circuit is at least substantially zero, said zero transmission frequency fo being located between f and fy in such a manner that the ratio:
 4. A flyback EHT and sawtooth current generator as claimed in claim 1, in which the transformer has a tertiary winding and in which the switching means and a tuning capacitor are connected to the primary winding, characterized in that the tertiary winding is connected through a parallel LC-circuit to the primary winding and that the network, at least due to the inductance of said coils, the capacitance of the tuning capacitor, the parasitic capacitance of the secondary winding and the rectifier circuit, the inductance and the capacitance of said parallel LC-circuit and the leakage inductance between the secondary and tertiary winding constitutes a network of a minimum 6th order with the said three resonant frequencies.
 5. A flyback EHT and sawtooth current generator as claimed in claim 1 in which a tuning capacitor is connected through a connection to the primary winding of the transformer, characterized in that a parallel LC-circuit is included in said connection and that the network, at least due to the inductance of said coils, the capacitance of the tuning capacitor, the parasitic capacitance of the secondary winding and the rectifier circuit, the inductance and the capacitance of said parallel LC-circuit and the leakage inductance between the primary and the secondary winding constitutes a network of a minimum of 6th order with the said three resonant frequencies.
 6. A flyback EHT and sawtooth current generator as claimed in claim 5, characterized in that the switching means are arranged across the series arrangement of the said tuning capacitor and the parallel LC-circuit. 